1. Field of the Invention
This invention relates to the field of self-tunable filters, and, in particular, to self-tunable continuous-time filters.
2. Background Art
In the field of electronic signal processing, continuous time (CT) filters have been used by designers to modify the phase and amplitude characteristics of analog signals. In the prior art, these filters have been constructed of discrete elements where each element has an accuracy within a desired range, or as an integrated circuit.
Integrated circuit continuous-time filters are often used for applications in the megahertz frequency range. Unlike switched capacitor filters, integrated circuit continuous time filters can avoid signal aliasing and noise problems. However, such filters have difficulty achieving equivalent tuning precision, stability and linearity. For example, in an integrated circuit CT filter, the pole frequency is inversely proportional to an RC (or g.sub.m /C) time constant, which is subject to large uncontrollable absolute component variations. Consequently, a post fabrication adjustment of the pole frequency to the nominal design value by means of trimming or tuning is generally required.
In the trim method, the pole frequency is adjusted only once, usually at wafer probe. The circuit is designed to minimize variations of the target cutoff frequency, Fc, over temperature and supply voltages as well as over the cutoff frequency range (in the case of a programmable filter). The advantages of this method are its simplicity and the absence of interference from a reference and on-chip tuning circuitry. The disadvantage is that it requires the measurement of Fc. Accurate measurement of a filter's pole frequency in the megahertz range is extremely difficult, especially on a production tester (e.g., Automatic Test Equipment, ATE). Further, wafer trimming is historically inaccurate and unreliable, and the device yields have been reduced, lowering the overall cost margins. The increased test time of the chips has also contributed to lower margins.
An alternative to trimming is the tuning method. The automatic or "self-tuning" of prior art continuous time integrated filters typically falls into one of two categories, a continuous tuning method whereby the filter can operate continuously while being tuned; and secondly, a method whereby the filter must be tuned in a signal-free environment. Continuous tuning methods have dominated prior art devices to date. An example of a continuous tuning system is illustrated in FIG. 3A, where the block diagram of a master-slave tuning control circuit is illustrated.
In the example of FIG. 3A, an input Vin is provided to main filter 301 through node 312. Main filter 301 outputs V.sub.out on node 324. V.sub.ref is provided at node 318, which is coupled to a master filter section 305, a frequency control block 303, and a Q control block 307. The master filter section 305 matches the performance of the main filter (slave) 301 normally used for signal processing. The master section may be a duplicate of the main filter or a subsection of the main filter which is sufficient to model its behavior. Tuning is performed by injecting a reference signal of known frequency at input V.sub.ref, comparing the performance of the main filter with that of the master, and then applying a correction signal to both the master and slave sections to eliminate the error.
Correction of frequency errors is achieved by the frequency control block which receives an input from master 305 through node 316 and detects frequency differences in the response of the master to the reference signal V.sub.ref to generate a control voltage at node 314 that is applied to master section 305 in a closed-loop fashion and to main filter 301. Master section 305 provides an output to Q control 307 through node 320. Since the behavior of the slave 301 is closely matched to that of master 305, the errors of the main filter 301 will be eliminated as master 305 is tuned in this fashion. Q control block 307 tunes the bandwidth of the filter by adjusting the pole quality factor Q in a manner similar to that of frequency control block 303. The error voltage for the Q tuning loop is generated by detecting the response of master section 305. The output of Q block 307 is provided to node 322, which is coupled to main filter 301 and to master section 305 in a closed-loop fashion.
For accurate tuning, the master-slave approach requires the matching of the master and main filter sections. Since filter circuits can occupy a large area in the integrated circuit design, master and main filter sections may be separated by some distance on the chip resulting in poor device matching. This is especially true of parasitic effects which can dominate high-frequency and high-Q designs, making it difficult to obtain good matching.
Due to the extra circuitry of the master filter section, the master-slave technique can also require excessive area in integrated circuit design such that the total area occupied by the filter is prohibitively large for chip designers. Depending on the configuration of the master section, the size of the tuning circuit may be 30-100% of the size of the main Filter.
Additional problems can arise when there is crosstalk between the master and slave sections. This is caused by the filter operating continuously while being tuned. The noise generated by the switching and control circuits of the tuner can also introduce inter-modulation distortion effects and degrade the overall signal-to-noise ratio of the system.
A second method of tuning in prior art devices is shown in FIG. 3B (tuning performed in a signal-free environment). FIG. 3B illustrates a biquadratic subsection of a filter. In the circuit of FIG. 3B, the biquadratic section is made to oscillate by introducing a negative resistance at the capacitive circuit nodes.
Referring to FIG. 3B, differential inputs Vin+ and Vin- are provided to a transconductance amplifier 207 through nodes 323 and 325 respectively. The positive output port of the transconductance amplifier 207 is taken at node 327, which is coupled to capacitor C7, the positive input port of buffer 311, and the negative output port of a transconductance amplifier 319. The negative output port of the transconductance amplifier 309 is taken at node 329, which is coupled to capacitor C8, the negative input port of buffer 311, and the positive output port of the transconductance amplifier 319. The positive and negative output ports of buffer 311 are coupled to the positive and negative input ports of transconductance amplifier 313, respectively, through nodes 331 and 333. The positive output port of a transconductance amplifier 313 is taken at node 335, which is coupled to capacitor C9, the positive input port of buffer 315, and the negative output port of a transconductance amplifier 317. The negative output port of the transconductance amplifier 313 is taken at node 337, which is coupled to capacitor C10, the negative input port of buffer 315, and the positive output port of the transconductance amplifier 317. Capacitors C7-C10 are coupled to a stable reference node 200, for example a ground. The positive output port of buffer 315 is coupled to node 339, which is coupled to the positive input ports of the transconductance amplifiers 317 and 319. The negative output port of buffer 315 is coupled to node 341, which is coupled to the negative input ports of transconductance amplifiers 317 and 319. Control voltage V.sub.g is fed to the transconductance amplifiers 309, 313, 317, and 319 through node 347 for tuning purposes. The positive and negative output ports of buffer 315 are connected to the lowpass output ports (Vout+ and Vout- ) of the second-order filter. A pair of transistors Q1 and Q2 are coupled to the biquadratic section. The transistor Q1 is coupled to the node 337 in its collector circuit. The transistor Q2 is coupled to the node 335 in its collector circuit. The emitters of Q1 and Q2 are coupled to node 349, which is coupled to current source 321. Current source 321 is coupled to node 351, which is coupled to switch ENOSC.
When ENOSC switch is closed, the negative impedance circuit, comprised of transistors Q and Q2, effectively causes g.sub.m2 to become negative, causing the biquadratic section to oscillate with a frequency near that of the pole locations. Once the oscillation frequency is detected, a correction signal is applied to trim the filter to its proper operating value. A disadvantage of this approach is the poor correlation of the oscillation frequency to the cutoff frequency. Due to parasitic effects and device mismatches in integrated circuit design, the oscillation frequency may not track the cutoff frequency of the filter with sufficient fidelity, producing poor tuning results.